Dielectric resonators are used in many circuits for concentrating electric fields. They are commonly used as filters in high frequency wireless communication systems, such as satellite and cellular communication applications. They can be used to form oscillators, triplexers and other circuits, in addition to filters. Combline filters are another well known type of circuit used in front-end transmit/receive filters and diplexers of communication systems such as Personal Communication System (PCS), and Global System for Mobile communications (GSM). The combline filters are configured to pass only certain frequency bands of electromagnetic waves as needed by the communication systems.
FIG. 1 is a perspective view of a typical dielectric resonator of the prior art. As can be seen, the resonator 10 is formed as a cylinder 12 of dielectric material with a circular, longitudinal through hole 14. FIG. 2A is a perspective view of a microwave dielectric resonator filter 20 of the prior art employing a plurality of dielectric resonators 10. The resonators 10 are arranged in the cavity 22 of a conductive enclosure 24. The conductive enclosure 24 typically is rectangular. The enclosure 24 commonly is formed of aluminum and is silver-plated, but other materials also are well known. The resonators 10 may be attached to the floor of the enclosure, such as by an adhesive, but also may be suspended above the floor of the enclosure by a low-loss dielectric support, such as a post or rod.
Microwave energy is introduced into the cavity by an input coupler 28 coupled to an input energy source through a conductive medium, such as a coaxial cable. That energy is electromagnetically coupled between the input coupler and the first dielectric resonator. Coupling may be electric, magnetic or both. Conductive separating walls 32 separate the resonators from each other and block (partially or wholly) coupling between physically adjacent resonators 10. Particularly, irises 30 in walls 32 control the coupling between adjacent resonators 10. Walls without irises generally prevent any coupling between adjacent resonators separated by those walls. Walls with irises allow some coupling between adjacent resonators separated by those walls. By way of example, the dielectric resonators 10 in FIG. 2 electromagnetically couple to each other sequentially, i.e., the energy from input coupler 28 couples into resonator 10a, resonator 10a couples with the sequentially next resonator 10b through iris 30a, resonator 10b couples with the sequentially next resonator 10c through iris 30b, and so on until the energy is coupled from the sequentially last resonator 10d to the output coupler 40. Wall 32a, which does not have an iris, prevents the field of resonator 10a from coupling with physically adjacent, but not sequentially adjacent, resonator 10d on the other side of the wall 32a. Dielectric resonator circuits are known in which cross coupling between non-sequentially adjacent resonators is desirable and is, therefore, allowed and/or caused to occur. However, cross-coupling is not illustrated in the exemplary dielectric resonator filter circuit shown in FIG. 2A.
An output coupler 40 is positioned adjacent the last resonator 10d to couple the microwave energy out of the filter 20. Signals also may be coupled into and out of a dielectric resonator circuit by other techniques, such as microstrips positioned on the bottom surface 44 of the enclosure 24 adjacent the resonators.
Generally, both the bandwidth and the center frequency of the filter must be set very precisely. Bandwidth is dictated by the coupling between the electrically adjacent dielectric resonators and, therefore, is primarily a function of (a) the spacing between the individual dielectric resonators 10 of the circuit and (b) the metal between the dielectric resonators (i.e., the size and shape of the housing 24, the walls 32 and the irises 30 in those walls, as well as any tuning screws placed between the dielectric resonators as discussed below). Frequency, on the other hand, is primarily a function of the characteristics of the individual dielectric resonators themselves, such as the size of the individual dielectric resonators and the metal adjacent the individual resonators (i.e., the housing and the tuning plates 42 discussed immediately below).
Initial frequency and bandwidth tuning of these circuits is done by selecting a particular size and shape for the housing and the spacing between the individual resonators. This is a very difficult process that is largely performed by those in the industry empirically by trial and error. Accordingly, it can be extremely laborious and costly. Particularly, each iteration of the trial and error process requires that the filter circuit be returned to a machine shop for re-machining of the cavity, irises, and/or tuning elements (e.g., tuning plates and tuning screws) to new dimensions. In addition, the tuning process involves very small and/or precise adjustments in the sizes and shapes of the housing, irises, tuning plates and cavity. Thus, the machining process itself is expensive and error-prone.
Furthermore, generally, a different housing design must be developed and manufactured for every circuit having a different frequency. Once the housing and initial design of the circuit is established, then it is often necessary or desirable to provide the capability to perform fine tuning of the frequency.
Furthermore, the walls within which the irises are formed, the tuning plates, and even the cavity all create losses to the system, decreasing the quality factor, Q, of the system and increasing the insertion loss of the system. Q essentially is an efficiency rating of the system and, more particularly, is the ratio of stored energy to lost energy in the system. The portions of the fields generated by the dielectric resonators that exist outside of the dielectric resonators touch all of the conductive components of the system, such as the enclosure 20, tuning plates 42, and internal walls 32 and 34, and inherently generate currents in those conductive elements. Field singularities exist at any sharp corners or edges of conductive components that exist in the electromagnetic fields of the filter. Any such singularities increase the insertion loss of the system, i.e., reduces the Q of the system. Thus, while the iris walls and tuning plates are necessary for tuning, they are the cause of loss of energy within the system.
In order to permit fine tuning of the frequency of such circuits after the basic design is developed, one or more metal tuning plates 42 may be attached to a top cover plate (the top cover plate is not shown in FIG. 2) generally coaxially with a corresponding resonator 10 to affect the field of the resonator (and particularly the parasitic capacitance experienced by the resonator) in order to help set the center frequency of the filter. Particularly, plate 42 may be mounted on a screw 43 passing through a threaded hole in the top cover plate (not shown) of enclosure 24. The screw may be rotated to vary the distance between the plate 42 and the resonator 10 to adjust the center frequency of the resonator.
This is a purely mechanical process that also tends to be performed by trial and error, i.e., by moving the tuning plates and then measuring the frequency of the circuit. This process also can be extremely laborious since each individual dielectric resonator and accompanying tuning plate must be individually adjusted and the resulting response measured.
Means also often are provided to fine tune the bandwidth of a dielectric resonator circuit after the basic design has been selected. Such mechanisms often comprise tuning screws positioned in the irises between the adjacent resonators to affect the coupling between the resonators. The tuning screws can be rotated within threaded holes in the housing to increase or decrease the amount of conductor (e.g., metal) between adjacent resonators in order to affect the capacitance between the two adjacent resonators and, therefore, the coupling therebetween.
A disadvantage of the use of tuning screws within the irises is that such a technique does not permit significant changes in coupling strength between the dielectric resonators. Tuning screws typically provide tunability of not much more than 1 or 2 percent change in bandwidth in a typical communication application, where the bandwidth of the signal is commonly about 1 percent of the carrier frequency. For example, it is not uncommon in a wireless communication system to have a 20 MHz bandwidth signal carried on a 2000 MHz carrier. It would be very difficult using tuning screws to adjust the bandwidth of the signal to much greater than 21 or 22 MHz.
As is well known in the art, dielectric resonators and dielectric resonator filters have multiple modes of electrical fields and magnetic fields concentrated at different center frequencies. A mode is a field configuration corresponding to a resonant frequency of the system as determined by Maxwell's equations. In a dielectric resonator, the fundamental resonant mode frequency, i.e., the lowest frequency, is normally the transverse electric field mode, TE01 (or TE hereinafter). Typically, the fundamental TE mode is the desired mode of the circuit or system in which the resonator is incorporated. The second-lowest-frequency mode typically is the hybrid mode, H11 (or H11 hereinafter). The H11 mode is excited from the dielectric resonator, but a considerable amount of electric field lies outside of the resonator and, therefore, is strongly affected by the cavity. The H11 mode is the result of an interaction of the dielectric resonator and the cavity within which it is positioned (i.e., the enclosure) and has two polarizations. The H11 mode field is orthogonal to the TE mode field. Some dielectric resonator circuits are designed so that the H11 mode is the fundamental mode. For instance, in dual mode filters, in which there are two signals at different frequencies, it is known to utilize the two polarizations of the H11 mode for the two signals.
There are additional higher order modes, including the TM01 mode, but they are rarely, if ever, used and essentially constitute interference. Typically, all of the modes other than the TE mode (or H11 mode in filters that utilize that mode) are undesired and constitute interference.
FIG. 2B is a perspective view of a conventional combline filter 100 (with a cover removed therefrom) having uniform resonator rods. As shown in FIG. 2B, the combline filter 100 includes a plurality of uniform resonator rods 106 disposed within a metal housing 102, input and output terminals 112 and 114 disposed on the outer surface of the metal housing 102, and loops 116 and 116 for inductively coupling electromagnetic signals to and from the input and output terminals 112 and 114. The metal housing 102 is provided with a plurality of cavities 104 separated by dividing walls 104a. Certain dividing walls 104a have a well-known structure called a decoupling “iris” 108 defining an opening in the wall. A dividing wall 104a a having an iris 108 is used to control the amount of coupling between two adjacent resonator rods 106, which controls the bandwidth of the filter. The resonator rods 106 vibrate or resonate at particular frequencies to filter or selectively pass certain frequencies of signals inductively applied thereto. Particularly, input signals from the input terminal 112 of the combline filter 100 are inductively transmitted to the first resonator rod 106 through the first loop 116 and are filtered through the resonance of the resonator rods 106. The filtered signals are then output at the output terminal 114 of the combline filter 100 through second the loop 116.
In conventional combline filters, the passing frequency range of the filter can be selectively varied by changing the lengths or dimensions of the resonator rods. The operational bandwidth of the filter is selectively varied by changing the electromagnetic (EM) coupling coefficients between the resonator rods. The EM coupling coefficient represents the strength of EM coupling between two adjacent resonator rods and equals the difference between the magnetic coupling coefficient and the electric coupling coefficient between the two resonator rods. The magnetic coupling coefficient represents the magnetic coupling strength between the two resonator rods, whereas the electric coupling coefficient represents the electric coupling strength between the two resonator rods. Usually, the magnetic coupling coefficient is larger than the electric coupling coefficient.
To vary the EM coupling (i.e., EM coupling coefficient) between two resonator rods, the size of the iris opening disposed between the two resonator rods is varied. For instance, if the iris disposed between the two resonator rods has a large opening, then a high EM coupling between the two resonator rods is effected. This results in a wide bandwidth operation of the filter. In contrast, if the iris has a small opening, a low EM coupling between the resonator rods is effected, resulting in a narrow bandwidth operation of the filter.
To vary the frequency of the filter, tuning screws (not shown in FIG. 2b) can be positioned so that they extend into the hollow center of the resonator rods. Such tuning screws can be adjustably mounted to the housing, such as by a threaded coupling, so that they can be screwed in and out so that more or less of the screws are disposed into the resonator rods. This alters the capacitive loading of the resonator rods and thus changes their center frequencies. This technique is shown and discussed in more detail in connection with FIG. 7 below.